The present invention relates to frequency discriminators, in general, and more particularly to a frequency discriminator for generating an output signal characteristically representative of a predetermined frequency spectrum of an input signal, but insensitive to variations in the amplitude and spectral width thereof, said frequency discriminator adaptable for use in a radar receiver clutter tracking loop to improve the filtering of clutter signals from the radar returns by maintaining a measured characteristic frequency of the clutter signal frequency spectrum substantially at a desired frequency with a loop response which is invariant to both amplitude and spectral width of the clutter signals.
Most modern airborne radars have the capability of detecting moving targets by distinguishing the moving target return signals from the clutter signals. Generally, this is accomplished by first suppressing or rejecting the clutter signals from the radar beam returns utilizing a clutter reject filter.
Problems associated with rejecting clutter return signals are particularly acute in airborne radars which scan their radar beam downward from a radar antenna mounted on a rotatable airborne antenna platform. Under these conditions, the radar beam transmitted signals not only echo from the moving target within the radar beam but also from any ground terrain background. A typical scenario is illustrated in FIG. 1. An airborne radar may be mounted on board an aircraft 20 having a radar antenna 22 which is capable of controlling an antenna beam .phi..sub..beta. thereof downward through a scanning or pointing angle .psi. referenced to a predetermined elevation level 24. The moving target for this example may be another aircraft like that shown at 26, for example, having a background of ground terrain 28. In the scenario illustrated, the aircraft 20 may be moving with a velocity v.sub.a and the aircraft 26 may be moving at a velocity v.sub.t.
For a pulse doppler airborne radar, the radar return signals during an interpulse period may appear as that shown by the graph of FIG. 2. In FIG. 2, the interpulse period falls between the pulsed transmissions at the times denoted as t.sub.0 and t.sub.1. The clutter return signal power denoted by the solid line 30, is generally much greater in magnitude than that of the return signal echoing from the target of interest which is denoted by the dashed line 32 occurring at approximately t.sub.t. As a result of the large differences in return signal magnitudes, it is generally very difficult to distinguish a target return signal from the clutter return signal without further processing.
For this reason, most doppler radars generally combine many successive interpluse periods to develop a frequency spectrum for each of a predetermined number of time increments or range cells of the interpulse periods in accordance with the return signal contents thereof. A graph depicting a developed doppler frequency spectrum for a range cell or group of range cells is exemplified in FIG. 3. Usually, the clutter return signal power, denoted by the solid line 34, is centered about a doppler centroid frequency f.sub.C. In this example, the clutter signal frequency spectrum 34 may be considered as having a bandwidth between the frequencies -f.sub.B and f.sub.B with respect to the doppler centroid frequency f.sub.C. On the other hand, the doppler frequency spectrums of typical moving targets may appear as that shown at 36 or 38 having doppler centroid frequencies of f.sub.t and -f.sub.t, respectively. As long as the doppler centroid frequency f.sub.C remains substantially fixed in the doppler frequency spectrum, a clutter reject filter may be tuned to the doppler centroid f.sub.C to filter out the clutter signals substantially within the bandwidth -f.sub.B to f.sub.B or thereabout. Accordingly, once the clutter signals are rejected from the frequency spectrum of the return signals, only the moving target signals remain for detection.
In most cases, however, the doppler frequency spectrum of the clutter returns is somewhat unstable particularly in the application of airborne radars where the doppler frequency spectrum of the clutter return varies with the aircraft motion in combination with the scanning of the radar antenna causing the doppler centroid frequency of the clutter signals to deviate from the tuned frequency f.sub.0 of the reject filter. To compensate for the variations of the centroid frequency of the clutter signals, some airborne radars have included a clutter tracking loop to maintain the doppler centroid frequency of the clutter signal frequency spectrum substantially at the tuned frequency of the clutter reject filter. A block diagram schematic of such a clutter tracking loop embodied within a typical radar is shown in FIG. 4.
Referring to FIG. 4, in a typical radar receiver, a transmitter 40 may generate transmitting signals which are conducted through a conventional circulator 42 and beamed over a spatial region via a rotatable antenna denoted at 44. Echo signals received from clutter and moving targets in the beamwidth of the radar are collected by the antenna 44 and conducted to an RF amplifier 46 via the circulator 42. Downstream of the RF amplifier 46 may be a plurality of mixers, a typical one or more of which being denoted at 48. The plurality of mixers are operative to convert the clutter and target return signal information from the RF portion of the spectrum through the IF down to the video or baseband portion of the frequency spectrum. Another set of mixers 49 and 50 may be included in the plurality to separate the clutter and target return signal information into in-phase (I) and quadrature (Q) components which may be thereafter conducted to a clutter reject filter and post processing apparatus (not shown).
The clutter tracking loop 52 generally includes an amplitude limiter 54, a frequency discriminator 56, an amplifier filter 58 and a voltage controlled crystal oscillator 60. Present clutter tracking loops generally use an analog split-filter discriminator or digitial signal frequency discriminator of the Foster-Seely type, for example, for implementing the function of the block 56. This type of frequency discriminator 56 is preceded by a hard amplitude limiter 54 so as to make the frequency discrimination operation carried out therein insensitive to the amplitudes of the received signal components I and Q. Both of the aforementioned discriminator types exhibit a characteristic steady-state transfer function which approaches the traditional "S" curve, such as that shown by the solid line 62 in the graph of FIG. 5. The gain (.delta.V/.delta.f), of the discriminator 56 (i.e. the slope of line 62) remains stable for the ideal case in which the clutter signal frequency spectrum is of a very narrow bandwidth.
In general, the predetermined or initial centroid frequency f.sub.C estimate for the clutter tracking loop 52 may be calculated from the aircraft navigation signals. This estimate denoted by the signal line 64, is input to the clutter tracking loop 52 via oscillator 60 for initializing the loop 52 during the acquisition operations thereof. The initial or acquisition frequency signal 64, in turn, governs the oscillator 60 to provide a frequency signal 61 to a selected one or more of the mixers 48 of the plurality to render the centroid frequency of the clutter spectrum at a predetermined baseband frequency which is generally zero frequency. In most modern air-craft, the navigation data is so accurate that the error between the actual and predicted centroid of the instantaneous clutter frequency spectrum is less than the PRF or sampling rate of the radar. This net error, however, is still too large to enable accurate target detection. Nonetheless, it is still possible to narrow the frequency discriminator bandwidth of the loop 52 such that return signals of interest like the moving target returns, for example, outside of the frequency range of the clutter frequency spectrum (see FIG. 3) are less likely to affect the loop 52 by pulling it off frequency lock.
However, a frequency discriminator with a frequency bandwidth only slightly wider than the maximum spectral width of the clutter signal frequency spectrum is characteristically subject to large changes in gain or slope as the spectral width of the clutter signal changes. For example, referring to the graph of FIG. 5, suppose that the initial or acquisition frequency f.sub.o of the loop 52 is set by the inertial navigation system via line In the ideal case, where the frequency spectrum of the clutter signal returns is very narrow, the frequency discriminator 56 may derive a frequency centroid f.sub.i thereof from the amplitude limited I and Q return signal components via unit 54 and in turn, generate an error voltage V.sub.1 (see curve 62 in FIG. 5) in response to a center frequency deviation (i.e., between the tuned frequency f.sub.o and centroid f.sub.i). The error signal V.sub.1 may be filtered and conditioned in the amplifier-filtering circuit 58 and passed along to the voltage control crystal oscillator 60 for governing the output frequency signal 61 thereof. Subsequently, the error signal V.sub.1 generated by the discriminator 56 is gradually caused to converge to zero as the instantaneous frequency centroid f.sub.i of the clutter frequency spectrum is controlled by the loop 52 to the tuned frequency f.sub.0 of the clutter reject filter (not shown). Under these ideal conditions, the response of the clutter tracking loop 52 is relatively stable and dependent on the gain (.delta.v/.delta.f).sub.1 or slope of the curve 62.
However, this is generally not the case in practice because the bandwidth of the clutter signal frequency spectrum is not ideally narrow and stable, nor is the frequency centroid thereof substantially fixed. For example, in the case of an airborne radar with a rotating antenna, the clutter signal spectral width is a function of the pointing angle .psi. of the scanning antenna and the aircraft velocity. Therefore, as the radar antenna is rotated through its scanning angle, the clutter signal bandwidth will vary responsively as a function of the scanned pointing angle .psi.. An example of this variation in clutter signal bandwidth for a forward sector scan of the radar antenna is illustrated in the graphs of FIGS. 6A and 6B. FIG. 6A exhibits the spectral width variation with respect to repetitious forward sector scans of the beam pointing angle .psi.. And accordingly, FIG. 6B illustrates the corresponding variation in the clutter signal spectral width relative to the forward sector scan of the antenna.
In response to these bandwidth variations of the clutter signal return, the gain of the frequency discriminator 56 is caused to change. For example, as the bandwidth of the clutter signal spreads out away from the ideal narrow case, the gain of the frequency discriminator 56 decreases as exhibited by the dashed line 70 in FIG. 5. With this reduced gain ##EQU1## a different error signal V.sub.2 may be generated by the discriminator 56 for the same instantaneous frequency centroid variation (f.sub.i -f.sub.o). As a result, the clutter tracking loop 52 takes a longer time in converging the instantaneous frequency centroid f.sub.i to the tuned frequency f.sub.0 of the clutter reject filter.
It is apparent then that as the clutter signal bandwidth varies, the response of the loop 52 will likewise vary proportionately therewith. Accordingly, this sensitivity to variation in the spectral bandwidth of the clutter signal promotes a less than stable response from the clutter tracking loop. In order to provide a fixed, stable response in the clutter tracking loop 52, a frequency discriminator having a transfer characteristic which is relatively independent of the amplitude and spectral width of the clutter return signals it is operating on is needed. It is the intent therefore in the instant application to disclose a more desirable frequency discriminator having characteristics improved over the discriminators presently used especially in the application to clutter tracking loops for stabilizing the loop response thereof.